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A view on electronics for the prototype of the GOSSIP detector in 0.13um CMOS Technology.

A view on electronics for the prototype of the GOSSIP detector in 0.13um CMOS Technology. Vladimir Gromov Electronics Technology NIKHEF, Amsterdam, the Netherlands December the 15 th , 2004. Highlights. Main functionalities of the detector and the principal block diagram of the detector.

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A view on electronics for the prototype of the GOSSIP detector in 0.13um CMOS Technology.

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  1. A view on electronics for the prototype of the GOSSIP detector in 0.13um CMOS Technology. Vladimir Gromov Electronics Technology NIKHEF, Amsterdam, the Netherlands December the 15th, 2004

  2. Highlights. Main functionalities of the detector and the principal block diagram of the detector. Main specifications of the electronics.(Single electron efficiency, time resolution, power consumption, analog-to-digital compatibility issue). A choice of sensitive pad-preamplifier coupling in the very front-end (calculation of the parasitic capacitances). Design of the preamplifier in the 0.13um CMOS technology (signal response, noise, hardness to the spread caused by the fabrication process instability). Design of the analog part of the read-out electronics in 0.13um CMOS. Performance of the detector featuring the design (efficiency, signal time-walk , overall time resolution). Block diagram of the DLL-based TDC. Current-steering logic is a way to eliminate switching noise in the mixed analog-digital design. Conclusion.

  3. The principal block diagram of the detector. • Main functionalities of the device: 1) The pixel structure with a fine pitch (30um …50um) can provide accurate information on X-Y coordinate of each cluster on the track. Thus the projection of the track is seen. 2) With having measured the drift time of each cluster the angle between the track and X-Y plane can be found in order to depict a 3D picture. • Design objectives on the read-out electronics: • ) the fact that the pixel has been hit needs to be detected with high efficiency and low faulty. The hit should be correctly related to a proper bunch-crossing. • ) drift time is to be measured as a latency of the hit arrival time in respect to the bunch-crossing signal accurate enough to determine Z - coordinate of the cluster. Cathode (drift) plane Track of the particle Cluster1 Cluster2 Z Cluster3 Drift distance Y Ingrid X Preamp Preamp Integral circuitin 0.13um CMOS technology Shaper Shaper Discriminator Discriminator Threshold Latch #1 Latch #1 4-bit DLL 1.6ns Latch #2 Latch #2 Latch #16 Latch #16 Clock 40MHz to Read-0ut

  4. Design objective the desirablevsthe possible Main specifications of the electronics: Single electron efficiency (input referred electronic noise). The fluctuations in the number of electrons in a single-electron avalanche is given by: P(n) = 1/M * exp(-n/M) , where M is a gas gain factor. With the input referred threshold at the level of 500e inefficiency will be 20% (gas gain M=2000) 10% (gas gain M=4000) 6% (gas gain M=8000) . The threshold of 448e corresponds to ENC = 90e RMS Inefficiency Gain=2000 Gain=4000 Gain=8000 20% 10% Threshold, electrons 448e

  5. 2) Time resolution. Both the time resolution of the TDC and the time-walk in the discriminator are independent contributors to the overall time resolution (σΣ ) of the electronics: σΣ = √(σ2TDS + σ2Time walk) , where σTDS = ∆t/√12 , ∆t – minimum bin size of the TDS σTime walk is dispersion, related to the time-walk in the discriminator For a 4-bit DLL-based TDC ∆t = 25ns/16 = 1.6ns it yields σTDS = 0.46ns . The electron drift velocity in the gas is about 20ns/mm therefore the TDC contribution to the overall spatial resolution will be σspatialTDS = 23um. Under these conditions time walk in the discriminator become the main contributor.

  6. Time-walk. Where does it come from? Signal at the output of discriminator (Sout(t)) is a convolution integral of input current i(t) and pulse response function (H(t)) of the electronics. Sout(t) Signals at the input of discriminator, arbitrary unit A fast shaped signal Low threshold A slow shaped signal High threshold Time, ns Range of time-walk for the fast shaped signal Range of time-walk for the slow shaped signal

  7. i(t) - input current : Ion current occurs in the Micromegas-pad gap in the period ∆tion= (∆L)2/μ U ≈ 30ns, where ∆L ≈ 50um is the Micromegas-pad distance U ≈ 400V is Micromegas-pad voltage μ= 1.72cm2V-1 sec-1 is mobility of ions in Argon Single-electron current in the detector Integral of current induced by a single electron. 0 i(t) 10% is electron contribution to the overall charge 0 90% is ion contribution to the overall charge ion component 0.5 0.05 S ( t1 ) s ( t ) electron component 1 0.1 1.5 0.15 20 0 20 40 20 0 20 40 time,ns time,ns t1 t

  8. H(t) - shaping function (δ-pulse response) . . . t Let us take shaping function of the electronics as follows F(p)=1/[(p (p +1)]. It demonstrates t + 1) 1 2 pulse response f (t, ) t , t 1 2 1 t t . , t , t f ( t 1 2 ) exp exp t t t t ( 1 2 ) 1 2 0.08 Pulse response of the electronics (τ1=1ns, τ2=10ns) 0.06 f ( t , 1 , 10 ) 0.04 f ( t , 8 , 10 ) Pulse response of the electronics (τ1=8ns, τ2=10ns) 0.02 0 0 10 20 30 40 50 60 t time,ns

  9. Distribution of the threshold-crossing time. Monte-Carlo simulations. Entries Gain=2000, Thr = 448e, τ1=1ns, τ2=10ns 600 400 Gain=2000, Thr = 448e, τ1=8ns, τ2=10ns 200 0 0 2 4 6 8 10 time,ns Integrals of the distributions. 4 . 1 10 Entries Gain=2000, Thr = 448e, τ1=1ns, τ2=10ns 5000 Gain=2000, Thr = 448e, τ1=8ns, τ2=10ns inefficiency=20% 0 0 2 4 6 8 10 time,ns !!! Fast shaping enables us to get much better time resolution at a given gas gain (threshold).

  10. 3) Power consumption. Number of channels per wafer = π D2/(4 pitch2) = 3.14 * 106 with power consumption 10W/wafer (possible to cool it down by gas flow) !!! Power consumption per channel = 3.2uW 4) Switching noise. In a mixed-mode design switching noise coming from digital part of the circuit back to high sensitive analog front-end is a very important issue. The most most common way to eliminate switching noise is using current-steering logic. Although it reduces speed and increase static power consumption. pitch =50um D=10cm

  11. The very front-end. DC or AC coupling to the Preamp. -800V R4 Cathode (drift) plane U1=-300V…-400V C4 R0 Ingrid C3 C0 ~ 20pF C2 C2 i(t) C1 R1 Preamp Preamp Preamp

  12. The very front-end. DC or AC coupling to the Preamp. Safety DC-coupling AC-coupling C5=C0*C1/(C0+C1) ≈ C1 C1 C0 Discharge trajectory Discharge trajectory C1 4*C2 Qd=U1*C0 4*C2 Qd=U1*C5 C3 C3 Zin≈0 R1 Zin≈0 Zin/4≈0 Zin/4≈0 C0 C0 Signal collection DC-coupling AC-coupling C1 i1(t) i1(t) iin(t) iin(t) 4*C2 C3 4*C2 C3 i(t) i(t) Zin≈0 Zin≈0 R1 Zin/4≈0 Zin/4≈0 C0 C0 In order to collect much of the charge iin(t) ≈ i(t) the following condition must be met C1 C3+4*C2. For better safety C1 0. Therefore values of the parasitic capacitors C3,C4 are important to know.

  13. Pad-to-Micromegas grid capacitance calculations. R - is a radius of the pad. The pad is a circle. d - is pad-to-Micromegas distance. - is vacuum dielectric constant. e 0 Ideal uniformly charged disk R Ideal boundless plane D C=1.8fF when R=25um, d=50um

  14. Pad-to-Pad capacitance calculations. b b a Ideal uniformly charged square pad Ideal uniformly charged square pad C=0.62fF when b=30um, a=20um, pitch=50um b - is length of the pad. The pad is a squire . 50um is a pitch [50um-b] - is a pad-to-pad distance. - is vacuum dielectric constant. e 0 - is relative permittivity of the dielectric. e r=4 Conclusion: C3+4*C2 = 1.8fF +4*0.62fF = 4.32fF In order to collect much of the charge iin(t) ≈ i(t) the following condition must be met C1 4fF

  15. Schematic of the Preamplifier. Main specifications. Technology: 0.13um CMOS. Power supply voltage:1.2V Power consumption: 1uA * 1.2V = 1.2uW. Charge sensitivity (real detector current pulse): 33mv/448e. Shaping function: rise time is 6ns, decay time is 100ns. Output noise (RMS): 4.3mV Equivalent input noise (RMS): (4.3mV/33mV)*448e = 58e . Idc=6nA T249 Id= -1u Gm = 2.4u Gds = 26n Vgs= -963mV Vds=-957mV (Vds_sat=-658mV) Cgg= 57.7fF Cdd+Cjd=0.451fF+0.150fF=0.6f 100MΩ Cdg=0.8fF Input Output T245 Id= 1u Gm = 23.2u Gds = 0.7u Vgs= 234mV Vds=243mV (Vds_sat=45mV) Cgg= 2.6fF Cdg=0.8fF Cdd+Cjd=0.8fF+0.7fF=1.5f

  16. The preamplifier. Simulation results. Spectral density of the squire of the noise output voltage |V2n(jw)| Input signal is a current δ-pulse Input signal is a real current pulse Output signal as a response to the δ-pulse Output signal as a response to the real current pulse 18.5uV2 4.3mV

  17. The preamplifier. Monte-Carlo analysis in Affirma Spectre. Channel-to-channel variations of output voltage Unom=247mV σ=6.5% Output Input Idc Channel-to-channel variations of the bias current Inom=1uA σ=5% Channel-to-channel variations of gain (charge sensitivity) GAINnom=33mV/448e σ=8.7%

  18. Schematic of the Preamp + Shaper + Discriminator. Main specifications. Technology: 0.13um CMOS. Power supply voltage:1.2V Power consumption: 1.6uA * 1.2V = 1.92uW (3.3*106 channels per wafer or 6.3W per wafer) . Charge sensitivity (real detector current pulse at the shaper output): 254mv/448e. Shaping function: rise time is 23ns, decay time is 100ns. Output noise (RMS): 37mV Equivalent input noise (RMS) at the shaper’s output: (37mV/254mV)*448e = 65e 300nA 1000nA 100nA 240nA Output Input

  19. Preamplifier + Shaper + Discriminator. Simulation results and Monte-Carlo analysis in Affirma Spectre. Channel-to-channel variations of the voltage at the output of the shaper σ = 18mV Uthr=190mV. Input current Preamp output Shaper output + threshold Channel-to-channel gain variations at the output of the shaper GAINnom=254mV/448e σ=10% Output of the first stage of the discriminator Output of the discriminator

  20. Preamplifier + Shaper + Discriminator. Statistical analysis. Walk-time as a function of the signal amplitude (THR=448e) 0.8*448e 50*448e 20*448e 12*448e 5*448e 3*448e 2*448e 1.5*448e 1*448e

  21. Preamplifier + Shaper + Discriminator. Statistical analysis. Time resolution Time-walk vs pulse height distribution Gain=8000 Gain=4000 Gain=2000 Time-walk curve Signal, electrons THR=448e Time resolution (time distribution of the threshold crossing events). Statistics is 10000. Entries Gain = 8000 efficiency=100% Gain = 4000 Gain = 2000 time,ns !!! It is feasible to reach time resolution of order of σ=2ns (100um) with a realistic gas gain.

  22. Block diagram of the DLL. Delay chain. DLL Clk 40MHz Phase detector #1 #2 #16 #3 #4

  23. An inverter in low-voltage current-steering logic. Vdd=1.2V bias1 Out- In + bias2 ±200mV Out+ In - bias3

  24. Conclusion. .The TimePix detector is going to be a powerful tool for future experiments. . Definition of the topology and specifications of the detector is in progress on the basis of the potentialities of the modern deep sub-micron CMOS technology . The following specification have been found feasible so far: Gas gain: 2000-8000. Single electron efficiency: 80%-94%. Input referred threshold: 500e. Time resolution: σ = 2ns corresponding to spatial resolution σ = 100um. Power dissipation: 3.2uW/channel (10W/wafer). AC coupling to the preamplifier looks preferable from safety point of view. Not much of the signal will be lost if the coupling capacitor is as tiny as 30fF…40fF. .First trial to design an analog circuit in the 0.13um CMOS technology capable to meet the specification has shown a promising result. . DLL-based TDC structure is a possible candidate for time-to-digital conversion block. . More efforts needs to be made to design switching noise free logic cells.

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