Digital Integrated Circuits A Design Perspective

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Digital Integrated Circuits A Design Perspective. Jan M. Rabaey Anantha Chandrakasan Borivoje Nikoli ć. Designing Combinational Logic Circuits. November 2002. Combinational vs. Sequential Logic. Combinational. Sequential. Output =. (. ). f. In, Previous In. Output =. (. ). f. In.

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Digital Integrated CircuitsA Design Perspective

Jan M. Rabaey

Anantha Chandrakasan

Borivoje Nikolić

Designing CombinationalLogic Circuits

November 2002.

Combinational vs. Sequential Logic

Combinational

Sequential

Output =

(

)

f

In, Previous In

Output =

(

)

f

In

At every point in time (except during the switching

transients) each gate output is connected to either

V

or

V

via a low-resistive path.

DD

ss

The outputs of the gates assumeat all timesthevalue

of the Boolean function, implemented by the circuit

(ignoring, once again, the transient effects during

switching periods).

This is in contrast to the

dynamic

circuit class, which

relies on temporary storage of signal values on the

capacitance of high impedance circuit nodes.

Static CMOS Circuit
Static Complementary CMOS

VDD

In1

PMOS only

In2

PUN

InN

F(In1,In2,…InN)

In1

In2

PDN

NMOS only

InN

PUN and PDN are dual logic networks

NMOS Transistors in Series/Parallel Connection
• Transistors can be thought as a switch controlled by its gate signal
• NMOS switch closes when switch control input is high
CL

CL

CL

CL

Threshold Drops

VDD

VDD

PUN

S

D

VDD

D

S

0  VDD

0  VDD - VTn

VGS

PDN

VDD 0

VDD |VTp|

VGS

D

S

VDD

S

D

B

A

C

D

Complex CMOS Gate

OUT = D + A • (B + C)

A

D

B

C

Properties of Complementary CMOS Gates Snapshot

High noise margins

:

V

and

V

are at

V

and

GND

, respectively.

OH

OL

DD

No static power consumption

:

There never exists a direct path between

V

and

DD

V

(

GND

.

SS

Comparable rise and fall times:

(under appropriate sizing conditions)

CMOS Properties
• Full rail-to-rail swing; high noise margins
• Logic levels not dependent upon the relative device sizes; ratioless
• Always a path to Vdd or Gnd in steady state; low output impedance
• Extremely high input resistance; nearly zero steady-state input current
• No direct path steady state between power and ground; no static power dissipation
• Propagation delay function of load capacitance and resistance of transistors
Rp

Rp

Rp

Rp

Rp

Rp

A

A

A

B

B

A

Cint

Rn

CL

CL

CL

Rn

Rn

Rn

Rn

B

A

B

A

A

Cint

Switch Delay Model

Req

A

A

NOR2

INV

NAND2

Rp

Rp

B

A

Cint

CL

Rn

A

Input Pattern Effects on Delay
• Delay is dependent on the pattern of inputs
• Low to high transition
• both inputs go low
• delay is 0.69 Rp/2 CL
• one input goes low
• delay is 0.69 Rp CL
• High to low transition
• both inputs go high
• delay is 0.69 2Rn CL

Rn

B

Delay Dependence on Input Patterns

A=B=10

A=1 0, B=1

A=1, B=10

Voltage [V]

time [ps]

NMOS = 0.5m/0.25 m

PMOS = 0.75m/0.25 m

CL = 100 fF

Rp

Rp

2

2

Rp

Rp

B

A

B

A

Rn

Cint

Cint

CL

CL

Rn

Rn

Rn

B

A

B

A

1

1

Transistor Sizing

4

4

2

2

A

D

Transistor Sizing a Complex CMOS Gate

B

8

6

4

3

C

8

6

4

6

OUT = D + A • (B + C)

A

2

D

1

B

2

C

2

D

C

B

A

C3

C2

C1

CL

Fan-In Considerations

A

Distributed RC model

(Elmore delay)

tpHL = 0.69 Reqn(C1+2C2+3C3+4CL)

Propagation delay deteriorates rapidly as a function of fan-in – quadratically in the worst case.

B

C

D

tpHL

tp

linear

tp as a Function of Fan-In

Gates with a fan-in greater than 4 should be avoided.

tp (psec)

tpLH

fan-in

tp as a Function of Fan-Out

All gates have the same drive current.

tpNOR2

tpNAND2

tpINV

tp (psec)

Slope is a function of “driving strength”

eff. fan-out

tp as a Function of Fan-In and Fan-Out
• Fan-in: quadratic due to increasing resistance and capacitance
• Fan-out: each additional fan-out gate adds two gate capacitances to CL

tp = a1FI + a2FI2 + a3FO

C3

C2

C1

CL

Fast Complex Gates:Design Technique 1
• Transistor sizing
• as long as fan-out capacitance dominates
• Progressive sizing

Distributed RC line

M1 > M2 > M3 > … > MN

(the fet closest to the

output is the smallest)

InN

MN

In3

M3

In2

M2

Can reduce delay by more than 20%; decreasing gains as technology shrinks

In1

M1

C2

C1

C1

C2

CL

CL

Fast Complex Gates:Design Technique 2
• Transistor ordering

critical path

critical path

01

charged

charged

1

In1

In3

M3

M3

1

1

In2

In2

M2

discharged

M2

charged

1

In3

discharged

In1

charged

M1

M1

01

delay determined by time to discharge CL, C1 and C2

delay determined by time to discharge CL

Fast Complex Gates:Design Technique 3
• Alternative logic structures

F = ABCDEFGH

CL

CL

Fast Complex Gates:Design Technique 4
• Isolating fan-in from fan-out using buffer insertion
Fast Complex Gates:Design Technique 5
• Reducing the voltage swing
• linear reduction in delay
• also reduces power consumption
• But the following gate is much slower!
• Or requires use of “sense amplifiers” on the receiving end to restore the signal level (memory design)

tpHL= 0.69 (3/4 (CL VDD)/ IDSATn )

= 0.69 (3/4 (CL Vswing)/ IDSATn )

Sizing Logic Paths for Speed
• Frequently, input capacitance of a logic path is constrained
• Logic also has to drive some capacitance
• Example: ALU load in an Intel’s microprocessor is 0.5pF
• How do we size the ALU datapath to achieve maximum speed?
• We have already solved this for the inverter chain – can we generalize it for any type of logic?
Buffer Example

In

Out

CL

1

2

N

(in units of tinv)

For given N: Ci+1/Ci = Ci/Ci-1

To find N: Ci+1/Ci ~ 4

How to generalize this to any logic path?

Ratioed Logic

Pseudo-NMOS VTC

3.0

2.5

W/L

= 4

2.0

p

1.5

[V]

W/L

= 2

t

u

p

o

V

1.0

W/L

= 0.5

W/L

= 1

p

p

0.5

W/L

= 0.25

p

0.0

0.0

0.5

1.0

1.5

2.0

2.5

V

[V]

in

V

V

DD

DD

M1

M2

Out

Out

A

A

PDN1

PDN2

B

B

V

V

SS

SS

Differential Cascode Voltage Switch Logic (DCVSL)

2.5

1.5

0.5

-0.5

0

0.2

0.4

0.6

0.8

1.0

DCVSL Transient Response

A B

[V]

e

A B

g

a

t

l

o

A

,

B

V

A,B

Time [ns]

Pass-TransistorLogic

NMOS-Only Logic

3.0

In

Out

2.0

[V]

x

e

g

a

t

l

o

V

1.0

0.0

0

0.5

1

1.5

2

Time [ns]

NMOS-only Switch

V

C =

2.5 V

C =

2.5

M

2

A =

2.5 V

B

A =

2.5 V

M

n

B

M

C

1

L

V

does not pull up to 2.5V, but 2.5V -

V

TN

B

Threshold voltage loss causes

static power consumption

NMOS has higher threshold than PMOS (body effect)

NMOS Only Logic: Level Restoring Transistor

V

DD

V

DD

Level Restorer

M

r

B

M

2

X

M

A

Out

n

M

1

• Restorer adds capacitance, takes away pull down current at X

• Ratio problem

2.0

1.0

0.0

0

100

200

300

400

500

Restorer Sizing

3.0

• Upper limit on restorer size
• Pass-transistor pull-downcan have several transistors in stack

W

/

L

=1.75/0.25

[V]

r

e

W

/

L

=1.50/0.25

g

r

a

t

l

o

V

W

/

L

=1.25/0.25

W

/

L

=1.0/0.25

r

r

Time [ps]

Solution 2: Single Transistor Pass Gate with VT=0

V

DD

V

DD

0V

2.5V

Out

0V

V

DD

2.5V

WATCH OUT FOR LEAKAGE CURRENTS

Solution 3: Transmission Gate

C

C

A

A

B

B

C

C

C =

2.5 V

A =

2.5 V

B

C

L

C

= 0 V

Transmission Gate XOR

B

B

M2

A

A

F

M1

M3/M4

B

B

2.5

2.5

2.5

2.5

V

V

V

V

V

V

1

i

n-1

i-1

i+1

n

In

C

C

C

C

C

0

0

0

0

(a)

R

R

R

R

eq

eq

eq

eq

V

V

V

V

V

1

n-1

i

i+1

n

In

C

C

C

C

C

(b)

R

R

R

eq

eq

eq

C

C

C

C

Delay in Transmission Gate Networks

m

R

R

R

eq

eq

eq

In

C

C

C

C

(c)

Similar delays for sum and carry

Dynamic Logic

Dynamic CMOS
• In static circuits at every point in time (except when switching) the output is connected to either GND or VDD via a low resistance path.
• fan-in of n requires 2n (n N-type + n P-type) devices
• Dynamic circuits rely on the temporary storage of signal values on the capacitance of high impedance nodes.
• requires on n + 2 (n+1 N-type + 1 P-type) transistors
Clk

Mp

((AB)+C)

Out

CL

A

C

B

Clk

Me

Dynamic Gate

off

Clk

Mp

on

1

Out

In1

In2

PDN

In3

Clk

Me

off

on

Two phase operation

Precharge (Clk = 0)

Evaluate (Clk = 1)

Conditions on Output
• Once the output of a dynamic gate is discharged, it cannot be charged again until the next precharge operation.
• Inputs to the gate can make at most one transition during evaluation.
• Output can be in the high impedance state during and after evaluation (PDN off), state is stored on CL
Properties of Dynamic Gates
• Logic function is implemented by the PDN only
• number of transistors is N + 2 (versus 2N for static complementary CMOS)
• Full swing outputs (VOL = GND and VOH = VDD)
• Non-ratioed - sizing of the devices does not affect the logic levels
• Faster switching speeds
• reduced load capacitance due to lower input capacitance (Cin)
• no Isc, so all the current provided by PDN goes into discharging CL
Properties of Dynamic Gates
• Overall power dissipation usually higher than static CMOS
• no static current path ever exists between VDD and GND (including Psc)
• no glitching
• higher transition probabilities
• PDN starts to work as soon as the input signals exceed VTn, so VM, VIH and VIL equal to VTn
• low noise margin (NML)
• Needs a precharge/evaluate clock
CLIssues in Dynamic Design 1: Charge Leakage

CLK

Clk

Mp

Out

A

Evaluate

VOut

Clk

Me

Precharge

Leakage sources

Dominant component is subthreshold current

CLSolution to Charge Leakage

Keeper

Clk

Mp

Mkp

A

Out

B

Clk

Me

Same approach as level restorer for pass-transistor logic

CL

CA

CB

Issues in Dynamic Design 2: Charge Sharing

Charge stored originally on CL is redistributed (shared) over CL and CA leading to reduced robustness

Clk

Mp

Out

A

B=0

Clk

Me

Cd=10fF

CL=50fF

Cb=15fF

Cc=15fF

Ca=15fF

Charge Sharing Example

Clk

Out

A

A

B

B

B

!B

C

C

Clk

Charge Sharing

V

DD

M

Clk

p

Out

C

L

M

A

a

X

C

a

M

=

B

0

b

C

b

M

Clk

e

Solution to Charge Redistribution

Clk

Clk

Mp

Mkp

Out

A

B

Clk

Me

Precharge internal nodes using a clock-driven transistor (at the cost of increased area and power)

CL1

CL2

Issues in Dynamic Design 3: Backgate Coupling

Clk

Mp

Out1

=1

Out2

=0

In

A=0

B=0

Clk

Me

Dynamic NAND

Static NAND

Backgate Coupling Effect

Out1

Voltage

Clk

Out2

In

Time, ns

CLIssues in Dynamic Design 4: Clock Feedthrough

Coupling between Out and Clk input of the precharge device due to the gate to drain capacitance. So voltage of Out can rise above VDD. The fast rising (and falling edges) of the clock couple to Out.

Clk

Mp

Out

A

B

Clk

Me

Clock Feedthrough

Clock feedthrough

Clk

Out

In1

In2

In3

In &

Clk

Voltage

In4

Out

Clk

Time, ns

Clock feedthrough

Other Effects
• Capacitive coupling
• Substrate coupling
• Minority charge injection
• Supply noise (ground bounce)
Clk

In

VTn

Out1

V

Out2

V

Clk

Clk

Mp

Mp

Out2

Out1

In

Clk

Clk

Me

Me

t

Only 0  1 transitions allowed at inputs!

Domino Logic

Clk

Mp

Mkp

Clk

Mp

Out1

Out2

1  1

1  0

0  0

0  1

In1

In4

PDN

In2

PDN

In5

In3

Clk

Me

Clk

Me

Ini

Ini

Ini

Ini

PDN

PDN

PDN

PDN

Inj

Inj

Inj

Inj

Why Domino?

Clk

Clk

Like falling dominos!

Properties of Domino Logic
• Only non-inverting logic can be implemented
• Very high speed
• static inverter can be skewed, only L-H transition
• Input capacitance reduced – smaller logical effort
np-CMOS

Clk

Me

Clk

Mp

Out1

1  1

1  0

In4

PUN

In1

In5

In2

PDN

0  0

0  1

In3

Out2

(to PDN)

Clk

Mp

Clk

Me

Only 0  1 transitions allowed at inputs of PDN Only 1  0 transitions allowed at inputs of PUN

NORA Logic

Clk

Me

Clk

Mp

Out1

1  1

1  0

In4

PUN

In1

In5

In2

PDN

0  0

0  1

In3

Out2

(to PDN)

Clk

Mp

Clk

Me

to other

PDN’s

to other

PUN’s

WARNING: Very sensitive to noise!